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by: D'angelo Will


D'angelo Will
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This 22 page Class Notes was uploaded by D'angelo Will on Thursday October 22, 2015. The Class Notes belongs to PORT 2 at University of California Santa Barbara taught by Staff in Fall. Since its upload, it has received 15 views. For similar materials see /class/226877/port-2-university-of-california-santa-barbara in Portuguese at University of California Santa Barbara.




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Date Created: 10/22/15
National Semiconductor Application Note OA 11 A Tutorial on Applying Op Amps to RF Applications September 1993 With operating frequencies exceeding 300MHz National s line of monolithic and hybrid current feedback operational amplifiers have become an attractive option for the RF and IF design engineer Typical operational amplifier specifications do not however include many of the common specifications familiar to RF engineers To help the designer exploit the many advantages these amplifiers can offer this application note will define the RF specifications of most interest to designers detail what determines each of these particular performance characteristics for National s current feedback op amps and where possible discuss performance optimization techniques To apply op amps to RF applications questions in three general areas must be addressed 1 Setting the op amp s operating conditions 2 Small signal AC performance in an RF context 3 Typical limits to RF amplifier dynamic range applied to op amps Wherever possible tested performance using the CLC404 will be used to demonstrate performance The CLC404 is a i5V power supply monolithic amplifier intended for use over a voltage gain range of 1 to 10 At its optimum gain of 6 the CLC404 offers a DC to 175MHz frequency range while delivering 12dBm power into a 509 load while dissipating only 110mW quiescent power National offers a wide range of additional monolithic op amps as well as higher supply voltage and hence higher power output hybrid amplifiers The best ampli er for a particular application will depend upon the desired gain power output frequency range and dynamic range Operation of National Current Feedback Op Amps The current feedback op amp developed by National Semiconductor Corporation provides a very wideband DC coupled op amp that has the distinct advantage of being relatively gainbandwidth independent As with all op amps using a closed loop negative feedback structure the frequency response for the National op amps is set by the loop gain characteristics The key development of the National amplifiers is to decouple the signal gain from the loop gain part of the transfer function This decoupling allows the desired signal gain to be changed without radically impacting the frequency response If compared to voltage feedback amplifiers which are constrained to a gainbandwidth product operation the current feedback topology offers truly 1993 National Semiconductor Corporation Printed in the U SA impressive equivalent gainbandwidth products eg the CLC401 at a gain of 20 yields a flat response with a 3dB bandwidth of 150MHz To match this a voltage feedback op amp would require 20150MHz BGHz gainband width product Please refer to National application note OA13 for a description of the current feedback op amp topology and transfer function One of the big changes in going from a classical RF amplifier to using an op amp is the exceptional flexibility offered by the op amps The designer is now charged with setting up the proper operating conditions for the op amp defining the gain and determining the lO impedances with external components Op amps allow the designer the option of running either a noninverting or an inverting gain path For RF applications the 180 phase shift provided by the inverting mode is often incidental There are however advantages and disadvantages to each mode depending on the desired performance and both will be considered at each stage in this development Most of this discussion on applying op amps to RF appli cations applies to any type of op amp The unique advantages of the current feedback topology are its high er frequency capabilities and its intrinsically low distortion at low operating currents If not specifically stated as being unique to the current feedback topology the items considered here apply equally as well to a voltage feed back op amp As a starting point for describing op amps for RF appli cations it is useful to summarize some of the standard operating assumptions for typical RF amplifiers Although there are certainly exceptions to the typical conditions shown here RF ampli ers generally have 1 AC coupled input and output A DC voltage generally has little meaning in RF applications Input and output impedances nominally set to 509 AC over the frequency range of operation This is seldom a physical 509 resister but rather a combi nation of active element lO impedances along with passive matching networks Fixed signal gain operations over a certain band of frequencies Any particular RF amp is purchased to provide a particular gain and is not user adjustable A two decade range of operating frequencies seems typical N 00 httpwwwnationalcom 4 Single power supply operation Since both input and output are AC coupled bipolar power supplies balanced around ground are not needed The DC bias point is maintained internally with minimal user adjustment possible Figure 1A shows a typical RF amplifier connection while Figures 18 and 1C show an ideal op amp either current or voltage feedback connected for noninverting and inverting gains respectively V Power supply decoupling not shown PI is assumed to come from a 509 509 loa G10 log P l GE Power gain fBrpm input to Figure 1A Typical RF Amplifier Connection For the RF amplifier both input and output are AC cou pled while a single power supply biases the part through Rb LC chokes off the AC output signal from seeing the power supply as a load The RF amplifier signal gain is specified with the output driving a 509 load and is defined as 10log power gain The two ideal op amp circuits assume that the source is coming from a ground referenced zero impedance volt age source while their outputs are intended to act as ideal zero ohm output impedance voltage sources to a ground referenced load The noninverting configuration ideally presents an infinite input impedance a zero ohm output impedance and a voltage gain as shown in Figure 18 from the plus input to the output pin Vcc Power supply decoupling not shown Figure 1B Ideal NonInverting Op Amp httpwwwnationaLcom The ideal inverting op amp differs in several respects from the noninverting The output voltage is ideally 180 out of phase from the input which accounts for the signal inversion The op amp s input ideally presents a virtual ground while drawing minimal current for either voltage or current feedback op amps This leaves Rg as the ideal input impedance seen by the source while the voltage gain from the input of R9 to the output is simply RfR This signal inversion is usually of no conse quence in an RF application and most ofthis discussion will deal only with the magnitude of the inverting gain Power supply decoupling not shown 2o H V Figure 1C Ideal Inverting Op Amp When using op amps as RF amplifiers we must first satisfy the lO impedance matching requirements recast the gain from a voltage gain to a power gain in dB and possibly configure for operation from a single power supply Figures 2A and 28 show the op amps of Figures 18 and 1c set up to provide lO impedance matching with the resulting power gain equations but still using bipolar supplies The bipolar power supplies allow operation to be maintained all the way down to DC Single supply operation is possible and will be considered next For the noninverting case setting Zi 509 simply requires a 509 termination resistor to ground on the non inverting input RT Getting ZO 509 simply requires a series 509 resistor in the output RO VE goal R 7 V G 20 log115 Log gain Given a desired 5 121o5quot Figure 2A NonInverting Op Amp Configured for RF Application For the inverting mode of op amp operation the input is ground referenced while the signal channel input impedance becomes the parallel combination of R and RM As 0A13 describes the current feedback topology depends on the value of the feedback resistor to determine the frequency response With each particular op amp calling out a particular optimum Rf R can then be used to set the gain and RM along with 9 will set the input impedance Setting R9 to yield the desired gain and then setting RM to satisfy Zi 509 will work until the required Rg lt 509 Having fixed Rf to satisfy the amplifier s stability requirements going to higher and higher inverting gains will eventually yield R s lt 509 Noninverting operation should be used ifthis imitation is reached Rf can however be increased beyond the recommended value for a current feedback op amp in order to allow an R 50 at higher gains but only at the expense of decreasing bandwidth 1R VL A 39ne lectin thesi nalinversion L 2R9 Vli 9 9 9 GT20log dB Loggain 2 R9 Given a desired GT Rf 2 109420 iRg Figure ZB Inverting Op Amp Configured for RF Application Note that for both topologies the gain to the matched load has been cut in half 6dB from the earlier ideal case through the voltage divider action of R0 RL It is a simple but critical conversion from any description of output voltage swing to and from a power in dBm defined at the load Figure 3 shows these conversions for a purely sinusoidal signal Basically for whatever initial description of voltage swing given we need to convert that into an RMS voltage square it and divide by the load RL 509 normally to get the absolute power in watts This is then divided by 0001 to reference that power to 1mW and 10log of that expression is taken to yield the power in dBm VLDD 2 ml Conversely for a given P0 in dBm 20log85090001 20log VLpp 4dBm VL 10P0 e420 DD Peak Peak voltage swing at load V0 2 10P0 e420 DD Peak Peak voltage swing at output pin Figure 3 Converting Between Voltage wings and Power Every op amp has a specified maximum output voltage swing that is generally shown as a peak excursion from ground This type of specification for balanced bipolar power supplies is really inferring how close the output may come to the supply voltages before nonlinear limit ing occurs For AC coupled RF applications it is always best to hold the output pin DC level centered between the two supply pins in order to provide the maximum output V Application note OA15 discusses in more detail input and output voltage range considerations Most of National s op amps do not require a ground reference for proper operation and can be easily operat ed from a single supply Generally all that is required is to keep the DC voltage on the input and the output pin centered between the voltages appearing on the two supply pins For a single supply operation with one supply pin held at ground this translates into the input and V0 being held at VCCZ For those amplifiers requiring a ground pin that pin should also be driven with a low source impedance voltage midway between the supply pins There are many possible implementations of single power supply op amp operation Figures 4a and 4b show two simple ways to operate noninverting and inverting op amps as AC coupled RF amplifiers using a single power supply In the noninverting case the input termination is still DC coupled while the input bias is set by the two Rb s to yield VCCZ Rb should be large enough to limit excessive quiescent current in the bias path but not so large as to generate excessive DC errors due to the amplifier s input bias current The gain setting resistor R9 is also AC coupled to limit the DC gain to 1 Hence the input DC bias voltage also appears at the output pin The output should be AC coupled in both circuits to limit the DC current that would be required if a grounded load were driven httpwwwnationalcom Ra V0 509 Figure 4a Single Supply Nonlnverting Amp Operation For the single supply inverting amplifier of Figure 48 we still require the midpoint reference to be brought in on the input A decoupling capacitor on that node is also suggested to decrease the AC source impedance for the noninverting input noise current The gain for this non inverting input reference voltage is again AC coupled to yield a unity DC gain to get VCCZ at the output pin The inverting input impedance goes from RM at DC to 509 at higherfrequencies RM as well as RT in Figure 4A could also be AC coupled to avoid DC loading on the source no 500 Figure 43 Single Supply lnverting Op Amp Operation For both of these single supply circuits we have given up the DC coupling for the signal path The low frequency limits to operation will now be set by the AC coupling capacitors along with impedances in each part of the circuit All of the subsequent discussions assume balanced bipolar supplies but apply equally as well to single supply operation Small Signal AC Performance Characteristics All of the typical small signal AC parameters specified for RF amplifiers are derived from the Sparameters reference 1 These are httpwwwnationalcom RF Amplifier Specifications Input VSWR Output VSWR Amplifier gain and bandwidth Reverse isolation Scattering Parameters S11 Input reflection 822 Output reflection 821 Forward transmission S12 Reverse transmission These frequency dependent specifications are measured using a network analyzer and an Sparameter test set A full 2port calibration should be performed prior to any device measurements The HP8753A used for the measurements reported here incorporates full 12 term error correction in its 2port calibration This basically normalizes all measurement errors due to imperfections in the cabling and test hardware reference 2 Figures 5A and 5B show the two configurations for the CLC404 used in demonstrating the small signal AC performance parameters listed above In each case the Sparameter test set places the device into a 509 input and output environment Both configurations achieve a voltage gain of 6 to the output pin and 3 to the 509 load This yields a gain of 20log3 954dB measured by the network analyzer Recall that one of the advantages to using op amps in RF applications is the exceptional flexibility in setting the gain Awide range of gains could have been selected for the test circuits of Figures 5A and 5B 6 was selected to allow easy comparisons to the CLC404 s data sheet specifications which are all defined at a gain of 6 Input Power Output Figure 5A Noninverting Amplifier Sparameter Test Circuit For the inverting gain configuration RM along with R9 sets the input impedance to 509 An RT of 509 is retained on the noninverting input to limit the possibility of selfoscillation in the noninverting input transistors See application note OA15 Figure SB lnverting Amplifier Sparameter Test Circuit InputOutput VSWR The Voltage Standing Wave Ratio VSWR is a measure of how well the input and output impedances are matched to the source impedance It is assumed throughout that the transmission line characteristic impedance is also equal to the source impedance of both portsSOQ in this case It is desirable that the input and output imped ances be as closely matched as possible to the source for maximum power transfer and minimum reflections Zr ZS VSWR or whichever gt1 Zs Z ZI gtamplifierinputoroutputimpedance Z S gt test system source impedance VSWR 1 VSWR 1 or 10log S 222 output Returnloss 20log J10logS112 input Ideal VSWR 1 Typically VSWR 15 for RFamps overtheiroperating frequency range Measuring the input VSWR is simply a matter of measur ing the ratio of the reflected power vs incident power on Port 1 of Figures SA and SB S11 A perfect match will reflect no power Output VSWR is measured similarly at Port 2 822 As described earlier an op amp s input and output impedances are determined by external components selected by the designer For this reason O VSWR is never shown on an op amp s data sheet Excellent VSWR can nevertheless be achieved using the compo nents shown in Figures 2A and 2B An op amp s gain polarity has minimal effect on the output VSWR At low frequencies RO by itself will determine the output VSWR Setting this resistor to 509 will yield excellent output VSWR to reasonably high frequencies As the test frequency increases however the op amp s output impedance will begin to increase as the loop gain rolls off reference 3 page 237 This induc tive characteristic can be partially compensated by a small shunt capacitance across RO Figure 6 shows this for either gain polarity along with tested output VSWR with and without this shunt capacitance The value of this capacitance will depend on the amplifier and to some extent on the gain setting and was determined empirically for this test by using a small adjustable cap 5 7 20pF directly across RO Mnsurlnq s Frequency MHz Figure 6 Measuring and Tuning CLC404 Output VSWR The marker at 200MHz indicates an output VSWR of 131 when CT is tuned optimally Tuning CT also extends the frequency response 821 slightly and will be left in place for the remainder of the tests The input impedance match of the noninverting topology Figure SA is principally set by RT As the frequency increases the input capacitance of the op amp will eventually degrade the input VSWR This effect is so negligible over the expected operating frequency range however that no tuning is required The input impedance match of the inverting topology Figure SB is at low frequencies set by the parallel combination of R9 and RM This holds very well as long as the amplifier s inverting input acts like a low imped ance over frequency For current feedback amplifiers the inverting input is actually a driven low impedance buffer It s impedance will however increase with frequency A voltage feedback amplifier s apparent inverting input impedance will also increase with frequency as its loop gain rolls off In the voltage feedback case the increase in inverting input impedance will be seen at a lower frequency than for a current feedback amplifier and will depend strongly on the amplifier gain setting Figure 7 shows the tested input VSWR for the two gain polarities of Figures SA and SB In this case we are measuring S11 and allowing the HP87SBA to convert the measurement and display VSWR directly httpwwwnationalcom Frequency MHz Figure 7 CLC404 Input VSWR Note carefully the change in scale for the input VSWR vs the output VSWR plot The marker on the noninverting test trace shows an exceptional input VSWR of 1031 at 200MHz while the inverting though higher remains under 1 41 through this range Forward Gain and Bandwidth Typical RF ampli er specifications show a fixed gain as defined in Figure 1A with a speci ed frequency range for 05dB gain flatness along with 3dB cutoff frequencies For the designer using a current feedback op amp a wide range of possible gains are easily obtainable Vl th the CLC404 s specified voltage gain range of 1 to 10 and including the additional 6dB loss from the output to the load 6dB to 14dB gains may be achieved using the CLC404 Higher gains can be achieved with this or any other current feedback ampli er with some sacrifice in bandwidth see application note OA13 For example the CLC401 specified over a 7 to 150 voltage gain range translates into an 11dB to 28dB gain range for RF applications The forward gain over frequency commonly called the frequency response and measured as 821 always appear in the National data sheets over a range of gains Small signal 3dB bandwidth and gain flatness are also guaranteed at a particular gain for each amplifier Rarely does a voltage feedback op amp show the 821 charac teristics since it so strongly depends upon the gain setting Rather these amplifiers show an open loop gain and phase plot and leave it to the designer to predict closed loop gain and phase The frequency response plots for the National op amps are normalized to show each gain coming in at the same grid on the plot for easier comparisons of frequency response shape over a wide range of gains Another advantage of the excellent loop gain control of the current feedback topology is exceptional forward gain phase linearity This phase is also shown on the frequency response plot A maximum deviation from linear phase is guaranteed at a particular gain setting in the data sheet speci cations The part to part variation in the frequency response is minimal forthe hybrid ampli ers from National with more httpmmnationalcom variation seen for the monolithic op amps As application note OA13 describes the current feedback topology allows an easy resistive trim for the frequency response shape that has no impact on the forward gain This frequency response flatness trim has the same effect for either noninverting or inverting topologies Figure 8 shows this adjustment added to the circuit of Figure 5A along with the measured 821 with and without this trim As OA13 describes this resistive trim inside the feed back loop has the effect of adjusting the loop gain and hence the frequency response without adjusting the signal gain which would still be set by only Rf and R This particulartest achieved a flatness of i1dB from D to 110MHz at a gain of 954dB for the noninverting test circuit shown with identical results for an inverting con figuration 954 dB 854 dB Frequency M Hz Figure 8 Measuring and Adjusting the Frequency Response 821 Note that the values for Rf and R have been reduced from those used in the circuit of Figure 5 although their ratio and hence the gain have remained the same Vl th the adjustment pot set to zero ohms this lower Rf value ensures that the frequency response will be peaked for any particular CLC404 used in the circuit Then by increasing the resistance into the inverting input the ampli er can be compensated and 821 adjusted to the excellent flatness shown above The part to part variation in frequency response becomes more pronounced as the desired operating frequencies and signal gains increase Operation of the CLC404 through 50MHz at 954dB gain would for example have minimal variation relative to operation through 100MHz and 14dB gain For i1dB atness and considering the rapid degradation in distortion performance at higher frequencies 100MHz is probably a reasonable upper limit for the operation of National op amps available at the time of publication in RF or IF applications Higher frequency operation can be achieved if the degraded flatness and distortion characteristics are acceptable to the application New product introductions can be expected to extend this operating frequency Reverse Isolation This small signal AC characteristic is a measure of how much signal injected into the output port makes it back into the input source The magnitude of S12 is the measure of reverse isolation National s current feedback op amps exhibit excellent reverse isolation relative to most RF amplifiers This results from both the output and inverting input being driven low impedance nodes To the extent that the output of the op amp and its inverting input both present very low impedances over wide frequency ranges significant signal attenuation can be expected in taking a signal voltage applied to the output matching resistor and tracing it back to either an inverting or noninverting input signal Slightly more attenuation can be expected for the noninverting vs inverting configurations since the signal must also get from the inverting to noninverting pin in the noninverting case The circuit of Figure 8 along with the inverting circuit of Figure 9a were used to measure the reverse isolation for both gain polarities as shown in Figure 9B Although reverse isolation is generally specified as a positive number this is simply the negative of the log gain in going backwards through the amplifier Hence the plot of Figure 9a shows a rising gain that would be interpreted as a decreasing reverse isolation as we go to higher frequencies As Figure 9B shows isolations in excess of 30dB are easily obtainable through frequencies far high er than the operating frequency range with very high isolations observed at low frequencies C Port 2 Figure 9A lnverting Reverse Isolation Test Circuit Frequan MHz Figure SB Reverse Gain for the Circuits of Figures 8A and 9A Dynamic Range Limiting Characteristics The final area of concern in applying op amps to RF applications are the limits to dynamic range familiar to RF amplifier users These are generally limited to ldB Compression Point 2Tone 3rd Order Intermodulation Intercept Noise Figure The ldB Compression Point is a measure of the maximum output power capability of the amplifier The 2Tone Intercept allows the prediction of spurious signals caused by amplifier nonlinearities when two input signals closely spaced in frequency are applied to the input The noise figure is a measure of how much noise is added by the amplifier and will set a limit to the minimum detectable signal Although each of these can certainly be measured for any particular op amp configuration their interpretation for op amps may vary from RF amplifiers depending on the op amp being used and the specification Each of these will be described generally and developed andor measured for the CLC404 with any anomalies in interpretation noted 1dB Compression Point Briefly stated this is the expected output power at a fixed input frequency where the amplifier s actual output power is 1dBm less than expected As Figure 10 shows it can also be interpreted as the ideal output power at which the actual amplifier gain has been reduced by 1dB from its value at lower output powers With both the X and Y axis of Figure 10 a dBm scale the output power vs the input power will have a slope of 1 If we shift the Xaxis by the amplifier s low power gain a 20dB gain was used arbitrarily in Figure 10 the amplifier s input to output transfer would ideally be a unity slope line through the origin An additional interpretation of Figure 10 is that beyond the ldB compression point the output power remains fixed as the input power is increased f 821 were measured at a fixed frequency with a swept input power we would get a horizontal line showing the low power gain that eventually transitions to a l slope line as the output power becomes fixed while the input power continues to increase The 1 dB compression power is commonly used as a maximum output power limit when computing an amplifi er s dynamic range Standard AC coupled RF amplifiers show a relatively constant 1 dB compression power over their operating frequency range httpwwwnationalcom Output Power dBm 51 8 3 8 o 10 Input Power dBm Figure 10 Illustration of 1dB Compression For an operational amplifier the maximum output power depends strongly on the input frequency The two op amp specifications that serve a similar purpose to 1dB compression are output voltage range and slew rate At low frequencies increasing the power of a fixed frequency input will eventually drive the output into the rails a saturation limit typically some number of diode drops below the supply voltages In addition as the input frequency increases all op amps will reach a limit on how fast the output can transition This is typically specified as a slew rate indicating the maximum dVdT at the output pin voltage Half this slew rate is available at the matched load when an output series matching resistor is used For a sinusoidal signal the maximum slew rate occurs at the 0 crossing This maximum dVdT is simply the peak voltage exursion times the radian frequency Given a slew rate in Voltssec SR and a frequency the maximum peak amplitude before slew limited operation is experienced is predicted to be SR2rrfrequency However this peak amplitude which can be converted to a dBm power at the load using the expressions devel oped earlier does not relate directly to the measured 1dB compression httpwwwnationaLcom Figure 11 shows the measured 1dB compression powers vs frequency for the CLC404 in the circuits of Figures 8 and 9 Since the maximum output power is principally a function of the output stage there is very little difference between the noninverting and inverting 1dB compression points For the National amplifiers that show a higher inverting slew rate than noninverting eg CLC400 a higher 1dB compression power at higher frequencies in inverting configurations would be expect ed The low frequency value however should be similar between polarities since it is determined by the maxi mum output voltage swing principally set by the power supply voltages and the headroom requirements in the output stage 1 B companion dBm Freqiancy 1 oMHzDiv Figure 11 1dB Compression for the CLC404 Although Figure 11 shows the 1 dB compression as defined in Figure 10 it is also very useful to look at the output waveforms and spectrums to gain an understanding of what is setting the measured 1 dB compression power Figures 12A and 128 show the time waveform and the spectrum at the load for the input power that yields the 1dB compression point for the CLC404 operating at 10MHz Load Vbllaue 1VIdiv Time Eonsdiv Figure 12A Output Waveform at 10MHz 1dB Compression om scam Frequency 10MHzdiv Figure 12b Output Spectrum at 10MHz 1dB Compression At this low frequency we are clearly running into an output voltage swing limitation With a 173dBm 1dB compression as shown at 10MHz in Figure 11 we would expect the fundamental amplitude in the spectrum to be at 163dBm The observed 16dBm in the spectrum of Figure 128 is a reasonable match to this expected fundamental power It is however incorrect to directly convert this fundamental power at 1dB compression into a sinusoid and expect that the amplifier can deliver a sinusoid of this amplitude For the 163dBm fundamental power predicted by the 1 dB compression measurement we might expect that the output is delivering an approxi mately 4Vpp sinusoidal swing at the load or i4V swing at the output pin Although this would exceed the maximum output swing specification for the CLC404 operating at 5 volt suppliesthis amplitude of sinusoid is in fact available if a zero loss filter is used to pass on only the fundamental harmonic Notice that a considerable portion of the output power has been spread into the odd order harmonics This is typical of the square wave output observed in the time domain trace of Figure 12A The fundamental 10MHz power can be related to the output time waveform ampli tude through the Fourier series expansion of the output waveform If the output were a perfect squarewave under conditions of output voltage limited operation a peak square wave amplitude of A would generate a fun damental frequency amplitude of 4Arr Going from the measured peak amplitude of the output time waveform the anticipated 1 dB compression would be calculated as the power in a sinusoid 41 times the square wave amplitude 1dBm Doing this for the measured i18V swing of Figure 12A would predict 151dBm peakpeak square wave amplitude converted to dBm 21dBm 20log4rr 1dBm reported 1dBm output power is 1dBm higher than measured power 183dBm This is 1dBm higher than measured This can be explained by the less than perfect square wave shape shown in the time waveform of Figure 12A This less than perfect square wave will yield a coefficient for the fundamental term in the Fourier expansion that is actually less than the predicted A4rr As the operating frequency increases the slew limit for the op amp will eventually restrict the achievable output swing to something less than the output voltage swing limit of the amplifier This can be observed in Figure 11 at approximately 30MHz for the CLC404 Again it is instructive to look at the time waveform and resulting spectrum when operating at an input power that yielded a 1dB compression in measured gain at the these higher frequencies Figures 13A and 138 show this for the noninverting circuit Figure 8 operating at the input power necessary to produce the measured 1dB compression with a 50MHz sinusoidal input signal From Figure 11 this input power would be 163dBm 954gain 68dBm Time Smdiv Figure 13A Measured Output Waveform at 50MHz 1dB Compression dam senm Fieqiency 50MHzdiv Figure 133 Measured Output Spectrum at 50MHz 1dB Compression The measured 1dB compression power under slew limited conditions is dependent on the amount of power in the fundamental frequency generated by the time waveform shown in Figure 13a Although we can say that the 1dB compression must be related to the amplifier s slew rate it would be very difficult to relate the slew rate to the waveform shape and then through the Fourier series to the fundamental power and hence 1dB compression The exact distribution of power into the fun damental and harmonics is changing over frequency All that can really be said is that at these higher frequency 1dB compressions a significantly distorted waveform httpwwwnationalcom with a peak to peak excursion less than that seen at lower frequencies is being generated At low frequencies the ldB compression power can be predicted approximately using the analysis shown earlier by assuming a square wave output set by the output voltage swing limits shown in the op amp data sheet Remember that the output voltage range specified in the data sheet is twice what can be delivered through the 6dB loss taken from the matching resistor to the load It is not however possible to easily predict the higher frequency ldB compression from the slew rate specification As will become apparent in the next section it is also not possible to relate the ldB compression to the third order intercept Typical RF amplifiers will show a 3rd order intercept 10dBm higher than the 1 dB compression point National op amps if they show an intercept characteristic have an intercept considerably higher than what would be predicted by adding 10dBm to the 1 dB compression 2Tone 3rd Order Intermodulation Intercept This specification is directed at predicting the 3rd order intermodulation distortion powers for any combination of two closely spaced in frequency input signals Any amplifier can be modeled to have a polynomial approxi mation to its transfer function from input to output When two input signal frequencies are present the 3rd order term of this polynomial approximation will give rise to distortion terms at frequencies that can be very near the input signal frequencies These closely spaced distortions are considerably more troublesome to narrowband IF channels than the simple harmonic distor tion terms that appear in integer increments away from the input signal frequency Appendix B expands all of the spurious frequency loca tions and distortion coefficients for two input signals at frequencies of foAf and foAf when passed through a 5th order polynomial With this simple definition of equal deviations from a center frequency an average frequency all of the spurious frequency locations become very simple algebraic expressions of f0 and Af Using this approach to defining the test frequency locations also allows a clear illustration of the symmetric clusters of spurious terms around integer multiples of f0 From appendix B the 3rd order intermodulation terms fall at fo r 3Afo With an input signal defined as Vi Acos2AfoAft Bcos2AfoAft and an input to out put voltage gain transfer function of V0 K0 K1 Vi K2Vi2 K3Vi3 ignoring the higher order terms for now a lower 3rd order spurious term at f0 3Af with an ampli tude of 34K3AB2 and an upper spurious at fo 3Af with an amplitude of 34K3A2B will result If equal amplitude signals were applied to the input A B and if these were increased in an equal fashion the two spurious amplitudes would increase in a cubic fashion In dBm terms if the two input and hence output powers httpwwwnationalcom were increased by 1dBm this model predicts that the two output third order spurious powers will increase by SdBm It is interesting to note the effect of adjusting just one of the input frequency power s Changing the lower test frequency power by 1dBm will change the lower spurious by 1dBm and the upper spurious by 2dBm Conversely changing the upper test frequency power by 1dBm will change the lower spurious by 2dBm and the upper by 1dBm The dependence of the 3rd order spurious power to output test frequency power assuming equal powers for each test frequency is shown graphically in Figure 14 40 30 Output and 3rd Order Spurious Powers dBm 239 20 30 Pa 3 Pa 40 P P rnw IoAI r rum I8AI IM3 E 3rd Order Interoepl P 3 20 0 10 30 40 Single Tone Input Power dBm Figure 14 Output and 3rd Order Spurious Power vs Input Power As shown in Figure 14 the 3rd order spurious powers increasing at a 3X rate vs the input power will at some output power intercept the desired output powers that are increasing at a 1X rate vs the input power Another way of saying this is that there is a 2X closure rate between the desired output powers and the undesired 3rd order intermodulation spurious powers The graph of Figure 14 was arbitrarily set up for an amplifier gain of 20dB the xaxis has been shifted to yield OdBm output power for 20dBm input power and for a 30dBm 3rd order intercept No actual amplifier will be able to reach the intercept point from an output power standpoint since this intercept typically exceeds the ldB compression power by at least 10dBm The intercept is intended as a mathematical construct to allow the prediction of the spurious power level for a given output signal power For an amplifier that shows a 3rd order intermodulation intercept characteristic a single measurement of output powers and spurious levels are sufficient to solve for the intercept point as shown by the equation in Figure 14 Figure 14 assumes equal output power levels for the two desired output signals A more general approach with unequal test power levels shows that from one set of measurements two estimates for the 3rd order intercept can be made Figure 15 steps through this analysis and concludes with the predictive equations for each of the two 3rd order spurious levels The graphical representa tion shown in Figure 15 is modeling what would be observed for a spectrum analyzer measurement of the test and spurious powers P P i 3M 3M 1 Al Af PL PH in st P2 7 PL 1 estimate ofintercepth31 P1 2 2nd estimate of intercept M32 P2 PEP Averaging these M3AVG g lfP1P2 PT and PL PH define AdBC Po iPL Po iPH IM3 Po Solving the two intercept estimates for PL and PH PL P2 72 IM37P1 PH P172 IM37P2 or for P1 P2 Po PL PH 2 SP0 4M3 Figure 15 3rd Order Intermodulation Intercept Calculations Typical RF amplifiers closely approximate this 3rd order intermodulation spurious model with an intercept that is relatively constant over the specified operating frequency range of the amplifier Op amps however show significant deviations from this simple model The principal difference is that all op amps will show a strongly frequency dependent 2tone 3rd order spurious performance The observed intermodulation spurious levels will be a function of the intrinsic distortion in the forward path of the amplifier corrected by whatever loop gain the amplifier has at that frequency All op amps show a loop gain that decreases with frequency Hence the 3rd order spurious levels will in general increase with frequency for fixed output test powers reference 4 discusses this loop gain dependence in detail An additional concern is at what point in the circuit to define the 3rd order intercept In order to make direct comparisons to RF amplifiers National defines the 3rd order intercept to be at a 50 load when driving from a 509 output impedance Some of the earlier National data sheets eg CLCZZO 0L0221 defined the intercept for a voltage swing at the output pin converted into a power as if it were driving 509 while actually applying that out put swing to the series 509 into a 509 load This has the effect of defining an intercept that is 6dBm higher than what is actually available at the load This can be seen from the equation shown above for IMS Recall that IMS for equal test power levels for the two test frequencies is simply the test power level 12 the difference between the test power levels and the spurious power levels This difference does not change in going from the output pin to the matched load However the output voltage swing will drop by 6dB and since the output pin power was erroneously defined as being a particular voltage swing across a 509 load when it in fact sees a 100 load this will translate into a 6dBm drop in the test power level to the matched 509 load Therefore the usable intercept at the matched load is 6dBm lower than specified in those earlier data sheets that call for an output power calculation at the output pin Given that the test power level is being defined at the matched load it is important to consider the amplifier limitations on the maximum power and frequency of test For a two tone test of equal powers and closely spaced frequencies the available peakto peak voltage swing for each test frequency at the load is 14 the peak to peak output voltage available at the amplifier s output pin while the available slew rate for each test tone can be estimat ed as 18 the amplifier s specified slew rate For a 2tone test signal being generated at a matched load twice the peak to peak swing is being generated in the envelope and twice the slew rate Going backthrough the match ing resistor to the output pin will double this swing and slew rate again In addition empirical testing has revealed that an overall maximum slew rate at the output pin that is 12 the specified op amp slew rate will show low spurious performance As the slew rate of the output pin waveform exceeds this limit additional non linearities come into play rapidly increasing the 3rd order spurious powers Using the circuit of Figure 8A and the typical specifica tions for the CLC404 the maximum test power level at the load for each test tone from an output swing stand point would be 146Vpp 15Vpp This translates into httpwwwnationalcom a maximum test power level for each tone of approxi mately 8dBm At this maximum output swing the available slew rate of 1 82000Vusec 250Vusec will limit the frequency of operation to less than SR2nVpp2 250E62n75V 53MHz As the test or operation powers decrease this upper frequency limit set by the slew rate limit will increase For example dropping the power 6dB to 2dBm will push this limit out to 106MHz Although some of the National current feedback amplifiers eg CLC400 CLC401 CLCSGO show a good approximation to the 3rd order intercept model the CLC404 used in the example circuits shown thus far shows a spurious power vs test power characteristic that deviates significantly from the simple model of Figure 14 Figure 16 shows the difference between the test and spu rious powers plotted as a function of single tone test power at the load Note that the independent variable axis is output power not the input power shown in Figure 14 Ideally this would at each frequency yield a straight line with a slope of 2 instead of the 3 slope shown in Figure 14 A similar plot for the CLC401 which more closely approximates this ideal is shown in Figure 17 If an op amp closely approximated the 3rd order intercept model a single measurement at one operating power would be adequate to predict the intercept at that frequency Spuious lavol dBc Figure 16 Measured 3rd Order Spurious for the CLC404 Spurious level dBc Single lune output power at 500 loan Figure 17 Measured 3rd Order Spurious for the CLC401 httpwwwnationaLcom The 3rd order spurious plot for the CLC404 is clearly showing some additional mechanism that is holding the spurious levels down as the output power level moves above OdBm At lower power levels it appears that the spurious characteristic is moving towards a linear slope of 2 as predicted by the simple intercept model Looking again at the 5th order expansion of the 2tone coefficients shown in appendix B an additional 5th order term contributes to the spurious powers observed at the 3rd order intermodulation frequencies Normally it would be expected that the K5 coefficient is so much lower than the K3 value that this 5th order contribution can be neglected However in the case of the CLC404 the K3 coefficient is so low as to make this second term significant at higher operating powers Note that the contribution of this 5th order term increases as the 5th power of the two test powers vs the more slowly increas ing 3rd order term It can be theorized that the 5th order coefficient is of opposite sign to the 3rd order coefficient Then as the test powers increase to the level that this 5th order term becomes significant in magnitude vs the 3rd order the spurious levels actually decrease for increasing output power The projected intercept at very low power levels can still be used to predict the spurious free dynamic range In Figure 16 the intercept at low output powers may be estimated for a particular frequency as the output power minus 12 the yaxis value However it should be realized that wideband op amps like the CLC404 actual ly provide better spurious performance at high powers than would be predicted by this low power intercept model The 3rd order intercept performance is typically very similar between inverting and noninverting topologies As discussed in reference 4 anything that changes the loop gain of the op amp will have an effect on the 3rd order spurious performance Increasing loop gain either by going to low feedback resistor values for current feedback op amps or low signal gains for voltage feedback op amps will decrease the spurious powers In both cases however increasing the loop gain by changing the external operating point is constrained by closed loop stability considerations 3rd order distortions and intermodulations can be further reduced by operating any 0 amp at higher quiescent currents if possible andor driving the output into a higher impedance load for those situations not requiring a 509 matched impedance environment Noise Figure Unlike the compression point and 3rd order intermodula tion intercept the noise figure for an op amp is always usable in the same way that it is for an RF amp It is important to remember that like compression and inter cept a noise figure is generally developed at a particular frequency and may change over frequency Normally however a single value can be used above the op amp s 1f noise corner frequency see application note OA12 for an additional noise discussion and its appendices for 1f noise corner discussion and tabulated op amp input noise terms for National op amps The noise figure can be accurately calculated from the equivalent input noise terms for an op amp and the resistor values used to achieve the desired gain and input impedance Unlike an RF amplifierwith a fixed gain and noise figure an op amp s noise figure will be strongly dependent on the gain setting We can however easily predict the noise figure with the equations developed here A very general development for an op amp s noninvert ing noise figure will be performed in order to allow easy comparison to noise figure expressions found in earlier National data sheets The inverting op amp s noise figure will however proceed with the assumption normally used that the input impedance is to be matched to the source impedance An idealized schematic illustrating the definition of noise figure is shown in Figure 18 Rs S So N 10log Si N N 80 Ni i o Noise Figure NF 10og i Figure 18 Noise Figure Definition All of the input and output noise and signal terms in the equation for noise figure NF are considered to be powers Ni is the noise power delivered by the source resistor to the input of the amplifier All other noise sources are considered to be part of the amplifier and contribute to the noise power No seen at the output Looking at the two parts of the NF expression inside the log function yields SiSO gt lnverse of the power gain provided by the amplifier NoNi gt Total output noise power including the contribution of RS divided by the noise power at the input due to RS To simplifythis consider Na as the noise power added by the amplifier reflected to its input port SiSO gt 1G NoNi gt GNiNaNi where GNiNa No Substituting these two expressions into the NF expression NF10og 10log1 The noise figure expression has simplified to depend only on the ratio of the noise power added by the amplifier at its input considering the source resistor to be in place but noiseless in getting Na to the noise power delivered by the source resistor considering all amplifier elements to be in place but noiseless in getting Ni Generally the definition for NF also constrains the input impedance for the amplifier to be conjugate matched to the source resistor this yields Ni k with this constraint We will however relax this constraint initially to allow comparison to the NF expressions found in National s earlier data sheets Eq 1 The NF of Equation 1 is specified in terms of a power ratio The individual noise terms for the op amp are how ever expressed as spot noise voltages or currents Spot means in a 1H2 bandwidth as opposed to integrated over some noise power bandwidth See OA12 Combining separately contributing noise sources is a matter of adding noise powers This can be done by converting all current noises to a voltage through the appropriate impedance then summing all of the squared noise voltage terms Any impedance normally needed to define a power or noise power bandwidth used to convert from spot to integrated noise will normalize out since we are developing the ratio of two powers at the same point in the circuitNa Ni Getting to the total spot noise power is then simply a matter of summing all the relevant squared noise voltages Figure 19 shows an op amp in the noninverting configuration with all of the individual resistor and ampli fier input noise terms detailed Where en op amp input voltage noise hi op amp noninverting input current noise i op amp inverting input current noise RT gt input terminating resistor eI RT s voltage noise ef Rf s voltage noise i9 Rg s current noise es Rs s voltage noise Figure 19 Noninverting Op Amp Noise Figure Analysis Circuit Recall thatthe noise ofa resistor Johnson Noise can be defined as either a spot current or voltage noise For a resister of value R these two possible expressions are voltage noise e R 4kTR 4kT current n0ise IR V httpwwwnationalcom where k gt Boltzman s constant k 138E23 Joules Kelvin T gt Kelvin 290 in this analysis 4kT 16E21 Joules at T 290 K The 3 amplifier noise terms are available for most of National s amplifiers in appendix 2 of OA12 If the spot noise Figure below the 1f noise corner is of interest appendix l of OA12 also shows how to approximate the low frequency spot noise from the high frequency flat band value and the 1f noise corner frequency Using the circuit of Figure 19 the NF expression can be developed by generating an expression for Ni and Na Ni is the noise power delivered by the source resistor noise to the input of the amplifier This analysis simply pro ceeds by considering the noise voltages as sources in normal linear circuit analysis but eventually squaring the resulting noise voltage delivered to RT from es Figure 2 shows the equivalent circuit and the resulting N This is considering the amplifier to have an infinite noninverting input impedance with all other noise sources neglected for now superposition of noise voltage contributions are used throughout this analysis Rs el RT RT e 4kTR 39 RSRT S 2 RT24kTRS Niei 2 RSRT RSRT define R R R p s T RSRT 4kTRpRT 4kTRp i RSRT 1RTRS Figure 20 Input Noise Power Calculation To get an expression for Na all other noise voltages and currents are referred to the noninverting input and summed as voltages squared For the noise terms on the inverting side ofthe amplifier it is best to find each term s gain to the output voltage then reflect back to the noninverting input by dividing by the noninverting voltage gain of the amplifier At this point since we are dealing with linear voltage gains define this gain as AV 1 RfRg Table 1 tabulates each individual voltage and current noise and its gain to the input of Figure 19 Note that all current noise terms have an impedance in their gain expression to yield all voltage noise terms at the input of the amplifier httpwwwnationalcom Voltage Noise Source Value Gain 0 Input Noninverting input 5 1 voltage noise n Noninverting input in RSIIRTgtRP current noise I lnverting input i R A current noise I f V Inputterminating I 4kTR9 RSRSRT re5istor voltage n0ise G 39 tt39 39 t cuill fnl ieres39m WTRg WV F 39 eedback resistor V4kTRg 1Av voltage n0ise Table I Noise Terms Contributing to Na for the Noninverting Op Amp Configuration One point of possible confusion is that although we are trying to develop the total noise power at the input of the op amp what relation does this have to the input voltage noise term that already appears in the op amp model en As described in OA12 the noise model for an op amp attempts to lump all the internal noise sources of the actual amplifier into an equivalent input noise voltage at the noninverting input and two input noise currents The intent is to provide a means of predicting the noise per formance over a wide range of external operating condi tions The en shown in the analysis model of Figure 19 is associated only with the internal characteristics of the op amp itself The total amplifier output noise includes this and contributions from all of the other noise sources shown there Having gotten to an expression for the total output noise voltage an equivalent input noise voltage may be derived by simply dividing by the voltage gain of the op amp This step of input referring each noise source is performed for each term in Table To form an expression for Na we need only to sum the squared product of each noise source and its associated gain as shown in Table l 2 2 2 2 2 R 4kTR R 4kTR 4kTR Naen 39niRP 39i T T 82 12 2 Av RstRT RgAv Av This will simplify to 4kTRp 4kTRf 1RT IRS AV 2 R Naen239niRp239i Av Eq 3 op amp noise terms input terminating combined feedback resistor noise term and gain setting resistor noise terms An expression for the noninverting noise Figure N may now be derived by substituting Equations 2 and 3 back into Equation 1 2 4kTR en2ime2i R 7 A 1RT IRS A NFT 10log 1 1km V 7quot HRSRT This will further simplify to HRSR HRSR R 2 em 2 iR 2 4kTR a i Eq4 Simplifying two of these terms RS RTRS 1 RT RT RS R T R R R 1T s T T Rs Rs R R RecallthatRp L RSRT R R R R R 1R Sjsz RT R SRT R then T T S T S 4kTRp 4kT 4kT Putting these simplifications back into Equation 4 yields 2 2 Rs em 2 mug H Eq 6 This expression for the noninverting noise Figure closely matches the equation shown in the CLCZOS and CLCZOG data sheets Equation 5 differs only in some of the vari able names and in the addition of a term due to the Rf and Rg noise which the CLCZOS and CLCZOG equations neglected iR RDA 4kTR AvtRp2 R NFquot 10log 1 5 RT If we were to let RT gt oo driving the signal directly into the noninverting input with no input termination and neglect any noise contribution from ini Rf and Rg Equation 5 will reduce to 2 eni2iinAV Eq6 NFT m10o 1 est 9 4kTRS This expression for NF matches that appearing in several of the National hybrid amplifier data sheets eg CLCZOO CLCZO l CLC lOS CLCZOS CLCZZO and CLCZZ where the Af term has been replaced by a1 in Equation 6 to consider only spot noise figure Equation 5 above is the most general expression for an op amp s noninverting spot noise figure considering an arbitrary input termination resister RT and all possible contributing terms even though some may prove negligible The simplified Equation 6 assumes no input terminating resister and neglects any noise contribution of the op amps noninverting input bias current noise and the feedback and gain setting resister noises The expression found in the CLCZOS and CLCZOG data sheets include an arbitrary RT but still neglected the noise contribution of Rf and R9 Having labored through all of this to clarify where some of the earlier noise figure expressions published by National came from we can now step to the most useful form of the noise figure expression where R5 RT Doing this in Equation 5 yields 2 2 R IR 4kTR eni2ini73 A39 A J NFT 10log 2 S With RS RT Eq7 This is probably the most useful formulation of the noise figure for the noninverting op amp The 2 arises from the signal attenuation we take in getting from the source to the input by using an external noisy resister matched to RS eg RT Note that the noise figure will decrease as the signal gain is increased due to the two numerator terms showing an AV in their denominators Also note that for current feedback amplifiers the feedback resister Rf is fixed to satisfy the amplifier s loop gain phase mar gin requirements application note OA13 discusses relaxing this requirement somewhat Hence the latter two terms in Equation 7 s numerator do indeed decrease with increasing gain lf Rf were not particularly constrained in value as with voltage feedback amplifiers the RfAV term appearing in the last two terms of Equation 7 would probably make more sense if replaced by an RfRg term Inverting Op Amp Noise Figure In this case the discussion will be simplified by constraining the input impedance of the op amp to be equal to RS Figure 21 shows the circuit for analysis with all of the contributing noise sources RT has been retained on the noninverting input along with its noise voltage source for complete generality Rg s noise now appears as a voltage source instead of the current noise term used in the noninverting analysis Again developing the noise figure expression for the inverting amplifier configuration is simply a matter of resolving Na and Ni and placing these expressions into Equation 1 Knowing that the input impedance is matched to RS 12 of the noise voltage attributed to RS will be delivered to the input port of the amplifier This yields a voltage 2 at the input httpwwwnationalcom Constrain RgIIRMRS AVT lnverting Gain ii R9 Figure 21 lnverting Op Amp Noise Figure Analysis 2 er2 ges Table II shows each individual noise terms except es with each term s gain to the inverting input The noise terms on the noninverting input have a gain of At to the inverting input This represents the noninverting gain to the output divided by the inverting gain back to the inverting input The two resistor noise terms for RM and R are taken to have a voltage gain to the inverting input defined simply by the resistor divider networks and simplified with the constraint on RM that it is to be set to yield RgRM R5 It is perhaps easiest to confirm the gain equations for R s and RM s noise by computing the current those voltages generate into Rg taking this current to the output by multiplying by Rf and then reflect ing back to the inverting input by dividing by AV RfRg Doing this and then substituting in for RM as shown In Table II will with some manipulation yield the simple gain expressions found in Table II The inverting noise current and Rf noise voltage are taken to the output then 1 Z4kTRS kTRS Eq 8 reflected back to the inverting input by dividing by the inverting gain Voltage Norse Source Value Glain tto npu Noninverting input en AT voltage norse Noninverting input ini RTAT curren no se Noninverting input Vm AT source resistor norse lnverting input i Rf current noise 39 AV lnverting input impedance 4leR 1 7amp matching resistor noise M 2 R9 Gain setting resistor 4kTR 171 voltage noise 9 2 R9 Feedback resistor 4leR 1 voltage noise f AV39 httpwwwnationaLcom Where 1R3 1AV 1 7 Rf 2Rg AV andAT Rg Ava 2Rf ForAV gt ATlt1 Rs R RQRS to etR R R mRQRS 9 M g 3 Table II Noise Terms Contributing to Na for the lnverting Op Amp Configuration The noise terms on the noninverting side of the op amp have a gain of At to the inverting input As Av increases this gain drops to lt l which contributes to the lower noise figure achievable using the inverting amplifier configuration Again an expression for a noise voltage 2 at the input may be obtained by taking the sum of the squared product of each noise source and associated gain shown in Table II 2 ea2enAT2 imRTAT2 4kTRTAT2 0 V 2 2 4kTRm 1 1 R S 4kTRg LB 4kTR 2 R9 2Rg f M2 Combining the two noise powers attributed to the input matching network will allow considerable simplification in the final inverting noise figure expression Substituting in for RM with the expression shown as part of Table II and expanding the squared gain expressions 2 R a 2 R 19 23 4kTRgl 27 5 4 R9 4 R9 R3 E9 rd4RgisRs 4kTRgRs RgiRs R3 R9 kTRs ri f kTRs 4 74 g S Putting this back into the inverting ea2 expression and grouping the noninverting input noise terms together yields 2 ea2 en2 iniRT2 4kTRTAT2 Alf kTRS4 v 4kTRf Adz Eq 9 R 973 Rs Putting the expressions for inverting ea2 and ei2 Equations 8 and 9 back into the noise figure expression Equation 1 and recognizing that dividing each term by RS will yield Na and Ni respectively shows that the kTRS term that arose in ea from the RM and R9 resistor noises will collapse to a simple term not Including kTRS very reminiscent of the 2 appearing in the NF expression One difference is that this part ofthe expres sion includes the contribution of both RM and Rg while the noninverting equation kept the RQ noise as part of the equivalent input noise This arises since R is now constrained by the input impedance matching require ment and can therefore be taken into this simplified form The inverting noise figure NF is then 2 R En2 imRT2 t 4kTRTAT2 ARL e 7 97 v v NF 710og 22 RS 1 kTRS Eq 10 To compare the noninverting noise figure expression Equation 7 to the inverting expressions Equation 10 note that noise terms on the noninverting input side have a gain of 1 for the noninverting configuration but a gain of At for the inverting Also note that the term associated with the feedback resistor noise is divided by simply AV in the noninverting case This arises because it also includes the RQ noise in the noninverting expression However it is divided by AV2 in the inverting case This arises from the R noise term being considered part of the input termination In this case Rg and RM s noise appears in 22R RS 1 as part of the noise figure expression Note at this collapses to simply equal 2 when R9 R8 similar to the NF case At low inverting gain the noninverting input noise terms have a larger impact for the inverting configuration than for an equivalent noninverting gain yielding a higher noise figure As Av increases however the noninvert ing noise terms will be attenuated in going to the invert ing signal input reference point yielding a lower inverting noise figure than for an equivalent gain noninverting configuration Figure 22 compares the noise figures over gain for the noninverting vs inverting configurations using the CLC404 For this comparison Rf is assumed fixed at 5009 RT for the noninverting case 509 but is set to 259 for the inverting case With these constraints R9 and RM will be set by the desired gain and the require ment that R RM R5 in the inverting mode R5 is assumed 09 throughout The inverting noise figure plot simply stops at the point where R 509 since higher gains are not possible with a gllixed Rf while retaining the input impedance matching requirement Also note that log gains are shown on the xaxis to the matched load while the voltage gains used in the noise figure calculations are the linear voltage gains to the output pin Noise Figure dB Figure 22 Noise Figure vs Gain For The CLC404 Dynamic Range Calculation Having developed the 3 limits to dynamic range commonly used in describing RF amplifiers as they apply to op amps it is now possible to combine them into a single dynamic range number The usable dynamic range is typically described in terms of the difference between the minimum detectable signal at the amplifier output and either the ldB compression or the output power that would bring the 3rd order spurious up to this minimally detectable level As described in reference 5 page 175 the minimally detectable signal at the output of an amplifier is POMDS kTGANF BX where kT gt Noise power delivered to matched input in dBm GA gt Power gain in dB NF gt Noise Figure B gt Noise bandwidth X gt Additional margin above the noise floor for detectability typically 3dBm Note that the kTGANF term solves to yield the spot output noise power Substituting for NF and recalling that kI noise power delivered to the input matching resistor N si Ni si kTGA NFNiGA W NOGA N0 Adding 10logB will then show the integrated noise floor at the output of the amplifier It is important to remember that this bandwidth need not be the bandwidth of the amplifier itself It is advantageous to bandlimit the response as narrowly as possible at some point after the amplifier immediately prior to the desired signal extrac tion It is this later bandlimiting bandwidth that would be used in the equation for determining the minimum detectable signal httpwwwnationalcom As an example consider the CLC404 circuit used throughout this discussion with a postfilter to yield a bandwidth of interest from 10MHz to 20MHz The non inverting gain of 954dB topology would yield a minimum detectable power level at the output POMDS 10log kT10log GA NF 10og 10MHz 3dB 7174dBm 954dB 18dB 70dB 3dB 7735dBm where the noise figure was read off of Figure 22 and kT 10log mji10og10004E7 21 7 7174dBm Having determined the minimally detectable signal at the output a maximum output signal set by some constraint will determine the dynamic range Typically a simple dynamic range specification uses the 1dB compression power as the maximum output power From the earlier discussion on 1dB compression we know that the actual output power at the fundamental frequency is 1dBm less than the reported 1dB compression point and that the true achievable sinusoidal power is approx imately 21dBm less than this due to increase in power showing in the fundamental when the output is approach ing a square wave With these considerations it would seem more realistic to use a maximum output power 3dBm less than the measured 1dB compression power Going to Figure 11 and subtracting SdBm from the mea sured 1dB compression at the maximum operating fre quency will yield a maximum useable output power atthe matched load of 145dBm Subtracting the minimum detectable signal at the output from this shows a 145 735 88dB dynamic range An alternative approach is to define a spurious free dynamic range This approach sets the maximum output power to yield a 3rd order spurious level equal to the min imum detectable signal At this point the amplifier gener ated spurious isjust equalling what can be detected from a noise floor consideration From Figure 15 the 3rd order spurious levels are PS 2 P0 4MB Setting this equal to the minimum detectable signal and solving for P3 2P0 7M3J7174dBm10ogGA 10ogB3dB 1 POmax 7174dBm10ogGA NF10ogB3dB2IM3 Putting in the previously develop minimum detectable signal and pulling the 20MHz intercept from Figure 16 shows a maximum spurious free output power of POW 735228 58 dBM httpwwwnationaLcom and a spurious free dynamic range of DRf 58dBm 735dBm 677dB An additional check on this spurious free dynamic range is to recall that the total output power for the equal power 2tone condition that will generate the spurious level at the minimum detectable signalis actually a voltage enve lope that is twice the individual signals or 6dBm higher in power This would imply a OdBm total output power when the spurious is just rising above the noise floor This is well below the dynamic range set by the 1dB compression limit Another way to interpret the 3rd order spurious plot of Figure 16 is to compute the absolute spurious power level as the output power is swept higher and simply compare that to the minimum detectable signal power at the output The absolute spurious power can be derived from the data of Figure 16 as simply the xaxis value minus the yaxis value POPOPSPS When the measured spurious free range is in a region of 1 slope 2 4dBm on the 20MHz line PS is remaining constant as the output power increases Using the actual mea sured data as opposed to an intercept can become a more appropriate way to compare the spurious power to the noise floorwhen the spurious level begins to level out and become constant for a part like the CLC404 just below the minimum detectable signal For example if the minimum detectable signal were actu ally 68dBm in the example considered earlier the 3rd order spurious would equal this level at 4dBm output power instead ofthe 4dBm level that would be predicted from the equation used above with the low power esti mate of a 28dBm intercept Increasing the amplifier gain by 6dB would raise the output noise floor to 675 to take advantage of this improved spurious performance at higher output powers Doing this would actually yield a 72dB spurious free dynamic range vs the 68dB calculated earlier The primary determinants to dynamic range are noise power bandwidth noise figure 1dB compression and 2 tone 3rd order spurious To maximize dynamic range the following steps may be taken 1 Limit the noise power bandwidth after the amplifier as much as possible To reduce noise figure run the amplifier at as high a gain as possible consistent with bandwidth limitations andor use the amplifier at high inverting gains Alternatively using a transformer coupled noninverting amplifier configuration as described in National application note OA14 can typically reduce the noise figure to the 6dB level If the 1 dB compression limits are inadequate use a higher supply voltage amplifier such the hybrid amplifiers offered by National or increase the power supply voltage above the recommended N F value National s low voltage monolithic amplifier s specify a maximum voltage across the supply pins of 14 volts Increasing the supplies from iSV to i6V or using a single 12V supply and the single supply circuits described earlier will typically increase the maximum useable output power by 2dBm Be sure to considerthe actual spurious performance if an intercept characteristic is not followed Increasing the supply current if possible or increasing the load impedance can dramatically drop the 3rd order distortion terms Recall that the feedback network remains as an upper limit on the output loading Reference 4 describes an additional technique of loop gain shaping that can be used to further improve the distortion performance 5 Conclusions High speed current feedback amplifier s can offer consid erable performance advantages when used in IF and RF applications The flexible gain and IO impedance capability can be used to the designers benefit in tailor ing the amplifier to the specific requirement Last minute gain changes can be accommodated with resistor value changes as opposed to requiring a new amplifier Exceptional IO VSWR and reverse isolation are easily attainable using wideband op amps Although somewhat different the dynamic range can be calculated or mea sured and compared between op amps and more typical RF amplifiers One of the most significant advantages of wideband current feedback amplifiers is the low 3rd order spurious level for their relatively low quiescent power dissipation Most of National s monolithic amplifier s dissipate less than 150mW while delivering in excess of 40dBm intercepts below 10MHz The primary drawback to the closed loop op amps are their rapid rolloff in distortion performance as the loop gain decreases at higher frequencies Another area for improvement are the relatively high noise figures using standard op amp topologies Using an input transformer can reduce the overall noise figure to around 6dB see National application note OA14 Additional external circuit techniques along with a new low noise op amp the CLC425 show a potential for noise figures as low as 2dB For IF and RF applications below 100MHz and particularly below 50MHz when a high spurious free dynamic range is required a wideband op amp solution can probably offer significant performance power dissipation and price advantages over more typical fixed gain amplifiers Appendix A summarizes the comparison between RF amplifiers and wideband op amps References 1 Hewlett Packard Application Note 154 SParameter Design Error Models for Systems Measurement Jim Fitzpatrick MicroWave Journal May 1978 Passive and Active Network Analysis and Synthesis Aram Budak Houghton Miflin 1974 Pushing Low Quiescent Power Op Amps to Greater than 55dBm 2Tone Intercept and an Automated Very Wide Dynamic Range System to Measure these Exceptionally Low Spurious Levels Michael Steffes National Application Note OA22 Microwave Transistor Amplifiers Analysis and Design Guillermo Gonzalez Prentice Hall Inc 1984 N F 5 01 httpwwwnationalcom Appendix A Amplifier Comparison Table Parameter RF Amplifiers National Semiconductor Op Amps Gain Bandwidth IO VSWR Reverse Isolation Noise Figure 3rd Order 1 dB Compression Supply Current httpwwwnationalcom Almost always xed gain Limited capacity lt 1MHz Can be very high frequency Generally 2 decade range Typically 151 20 to 30dB considered good Not too frequency dependent Can be very low 2 to 5dB typical Moderate levels needs high lcc for excellent numbers Relatively frequency independent Good levels for voltage supplies Relatively frequency independent 1dB 3rd intercept are order related Usually single polarity supply High quiescent current vs PoCapability 20 Easily adjustable over very wide range DC capability Upper limit around 100MHz to match RF amp flatness specs Can be tuned to much better match through 100MHz than RF amps Much better isolation possible Degrades at high frequencies Better noninverting than inverting Varies with gain setting Higher gains better but bottoming out about 12dB for typical op amp Circuit can be improved to lt 5dB Very good intercepts for quiescent power Strong frequency depen dentdegrading rapidly at high frequency Can be improved at low frequencies See App Note OA22 Requires more head room for avail ble output power Drops rapidly with frequency due to slew rate 1dB and 3rd order not related Bipolar supplies Almost all can be run single supply Much lower lower quiescent currents for PoCapability Appendix B Harmonic amp Intermodulation Terms for a 5th Order Polynomial Transfer Function For an input signal that is two sinusoidal signals Vi Acos27rf1t Bcos21rf1t f1 f0 Af f2 f0 Af Processed through a 5th order polynomial transfer function V0 K0 K1Vi K2Vi2 K3Vi3 K4Vi4 K5Vi5 yields the following frequencies and coefficients Frequency terms and coef cients in order of ascending frequency Frequency Coefficient 2 2 DC K0K2 A B K4 3A4 334 3932 2 2 a a 2 2Af K22K4 A33AB3 K5 A4BAB4 2nd Order Intermod 1o 3 2 f0 5Af K5 EA B 5th Order Intermod 3 2 3o 2 3 f0 3Af K3 IAB EK5A B 3rd Order Intermod foAffr K1K3A3 A2BJK5 A5A382AB4 LowerTestTonen f0 No coef cient fOAff2 K1K3 B3 AB2JK5 B5A283A4BJ UpperTestTonef2 f0 3M K3 AZBK5A382 3rd Order Intermod f0 5M K5 A283 5th Order Intermod 2f0 4Af 2K4A3B 4th Order Intermod A2 A4 3 2 2 2f0 2Af 2fr K2 TK4 T K4A B 2nd Harmonic for fr 2f0 K2 gum A3BAB3 K5A4BAB4 2nd Order Intermod 32 B4 3 2 2 2f0 2Af 2f2 K2 TK4 T K4A B 2nd Harmonic for f2 21 httpwwwnationalcom Appendix B Continued Harmonic amp Intermodulation Terms for a 5th Order Polynomial Transfer Function Frequency Coefficient 2f0 4Af 2K4A83 4th Order Intermod 3f0 5M K5A4B 5th Order Intermod 3f0 3Af 3n KTsAs f 2A5 3rd Harmonic for f1 3f0 Af K3ABZ K5AZB3 Higher 3rd Order Intermod 3f0 No Coef cient 3f0 Af K3AZBK5A382 Higher 3rd Order Intermod K3 K5 3f0 3M 3f2 T33 WB5 3rd Harmonic for f2 3f0 5Af K 5 AB 4 5th Order Intermod 4f0 4A 4g K4A4 4th Harmonic for f1 4f0 2Af 2K4 A3 B 4th Order Intermod 4f0 No Coef cient 4f0 2Af 2K4 A83 4th Order Intermod 4f0 4Af 4f2 gK B4 4th Harmonic for f2 5f0 5Af Sfr K5A5 5th Harmonic for fr 5f0 3M K5 A4 B 5th Order Intermod 5f0 No Coef cient 5f0 3Af K5 AB4 5th Order Intermod 5f0 5M 5f2 K5 B 5 5th Harmonic for f2 httpwwwnationalcom


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